The present invention pertains to an improvement of a composite power amplifier such as a Chireix power amplifier (See Non-patent Document 1) and a Doherty power amplifier (Non-patent document 2), the improvement directed to remove a disadvantageous narrow band characteristic of such composite power amplifier. The Chireix power amplifier was invented by Chireix in 1935, and the Doherty power amplifier was invented by Doherty in 1936. Both of them have been used as AM broadcasting transmitters. Before semiconductors were invented, their power amplifying elements were vacuum tubes. However, after semiconductor amplifiers were invented, they were widely used as final-stage amplifiers, especially in mobile, satellite, and broadcasting systems. The Chireix amplifiers are often called LINC power amplifiers (Non-patent Reference 3).
In the Chireix amplifiers and the Doherty amplifier (hereinafter referred to as composite amplifiers), the ratio between an output voltage and an output current, known as equivalent load impedance, is varied in accordance with the magnitude of an input signal, thereby increasing the power efficiency of an amplifier. This makes their amplification behaviors extremely complex, so that it is difficult to maintain a high power efficiency and a linearity over a wide frequency band.
An operational principle and a power efficiency characteristic of an prior art composite power amplifier will now be given below with reference to the Chireix amplifier first and then to the Doherty amplifier.
As shown in FIG. 1, a Chireix transmitter 101 equipped with the Chireix power amplifier comprises;
(1) a signal component separating device 190 which, upon receipt of an in-phase component I(t) of a baseband signal (the in-phase component I(t) hereinafter referred to as in-phase signal I(t)) and an orthogonal component Q(t) of the baseband signal (the orthogonal component hereinafter referred to as orthogonal signal Q(t)), generates
an RF modulated signal A (hereinafter referred to a main signal A), obtained by orthogonal modulation of I(t) and Q(t) (Eq. 1 below),
a signal B orthogonal to the main signal A (the orthogonal signal B hereinafter referred to as efficiency improving signal EIS B), and
a first composite signal S1 in the form of a vectorial sum of the main signal A and the EIS B, and
a second composite signal S2 in the form of a vectorial difference between the main signal A and the EIS B;
(2) power amplifiers 150 and 151 for power amplifying the respective composite signals S1 and S2; and
(3) a Chireix power combiner 140 for combining the outputs of the power amplifiers 150 and 151 to provide the output signal S0 of the Chireix transmitter 101.
The Chireix network 140 consists of two impedance inverters (or ¼-wavelength lines) 160 and 161 each adapted to interface its input terminals and its output terminal; and two reactance elements 170 and 171 each adapted to interface its respective input terminals and the ground. The reactances of the two reactance elements have the same absolute value, but have opposite signs. It is noted that the term “efficiency improving signal (EIS)” is not a common technical term but is a term named by the present inventor to connote an implication of the signal that it enhances the power efficiencies of the power amplifiers. This term will be used in this context in an exposition below of the prior art Doherty transmitter and in the description of the present invention as well.
The main signal A is a high-frequency modulated signal (or RF modulated signal) obtained by an orthogonal modulation of an in-phase component of a baseband signal (hereinafter referred to as in-phase baseband signal) I(t) and an orthogonal component of the baseband signal (hereinafter referred to as orthogonal baseband signal) Q(t) with a carrier angular frequency ω0. Thus, the main signal A is given by the following equation.A(t)=I(t)cos(ω0t)+Q(t)sin(ω0t)  (Eq. 1)
In terms of an envelope signal a(t) and a phase modulation signal φ(t), the main signal A and its vectorial form a can be written in the following forms.A(t)=a(t)cos {ω0t+φ(t)}  (Eq. 2)a=a(t)exp{jω(t)}  (Eq. 3)
An EIS B, which is orthogonal to the main signal A, satisfies a condition that the envelope of the composite signal of the main signal A and the EIS B equals the peak envelope level C (which is often referred to as peak level C or simply C) of the main signal A. From this condition, the EIS B is given by the following equation.B(t)=−b(t)sin {ω0t+φ(t)}  (Eq. 4)where b(t)=√{square root over (C2−a(t)2)} and C is the peak level of the main signal.
The power amplifiers 150 receives the first input signal S1 which is a vectorial sum of the EIS B (for which a vector representation is jb) and the main signal A, while the power amplifier 151 receives the second input signal S2 which is a vectorial difference between the main signal A and the EIS B, as given by the following Equations.S1(t)=A(t)+B(t)  (Eq. 5)S2(t)=A(t)−B(t)  (Eq. 6)
Plugging Eq. 2 and Eq. 4 in Eq. 5 and Eq. 6, respectively, S1(t) and S2(t) turn out to beS1(t)=C cos [ω0t+φ(t)+cos−1{a(t)/C}]  (Eq. 7)S2(t)=C cos [ω0t+φ(t)−cos−1{a(t)/C}]  (Eq8)
Eq. 7 and Eq. 8 show respectively that the envelope levels of the first and the second composite signals S1 and S2, respectively, are constant (equal to the peak level C of the main signal) and that they are either advanced or delayed in phase relative to the main signal by cos−1{a(t)/C}.
Since in the Chireix transmitter 101 the envelope levels of the input signals fed to the power amplifiers 150 and 151 are always equalized to the peak level C of the main signal by adding the EIS B to the main signal A, the power amplifiers 150 and 151 always operate at their maximum power efficiencies. The EISs B fed to the input ends of the power amplifiers 150 and 151 are cancelled out at the output end of the Chireix power combiner 140, resulting in an output signal S0 given by the following Eq. 9.S0(t)=g{S1(t)+S2(t)}/√{square root over (2)}=√{square root over (2)}gA(t)  (Eq. 9)where g is the voltage gain of the power amplifiers 150 and 151.
Focusing attention to the power amplifier 150, it appears that, because of the power amplifier 151 in operation, an equivalent output impedance (i.e. the ratio between the output voltage and the output current) of the power amplifier 150 turns out to be a non-real number. In other words, it appears that an element of reactance Xc is connected between the output end of the power amplifier 150 and the ground. In order to achieve an impedance matching with the power amplifier 151, it suffices to connect an reactance element 170 of reactance −Xc between the output end of the power amplifier 150 and the ground. In this case, however, the magnitude of Xc changes with the envelope level a(t) of the main signal A, so that the impedance matching can be achieved only when the main signal A has certain particular magnitudes. Referring to FIG. 2, there is shown a power efficiency characteristic (curve c21) of the Chireix transmitter 101, in which the abscissa represents the normalized input voltage of the main signal A and the ordinate the power efficiency. It is shown in the figure by curve c21 that the power amplifier 150 has a maximum power efficiency η0 only when a=a1 and a=a2 where a is the normalized input voltage of the main signal A, but has a power efficiency less than η0 otherwise (especially when a<a1) (Non-patent Document 4). It should be noted that curve c21 represents a case where the power amplifiers 150 and 151 are assumed to be B-class amplifiers and Xc is chosen to maximize the power efficiency of the Chirex transmitter 101 at a1=⅓. Curve c20 shows a power efficiency of the B-class amplifier. One way to improve the power efficiency of the Chireix transmitter 101 is to construct the transmitter in a power recycling configuration (Non-patent Document 5). FIG. 3 shows a power recycling type Chireix transmitter 102, in which, unlike the Chireix transmitter 101, the main signal A and the EIS B of the outputs of the power amplifiers 150 and 151 are extracted independently by means of a 180-degree hybrid circuit 141, and the power of the extracted EIS B is converted into a DC power by a high-frequency-DC converter circuit 172 and recycled to power supply terminals Vs of the power amplifiers 150 and 151. If the high-frequency-DC converter circuit 172 had 100% power conversion efficiency, then the power efficiency of the Chireix transmitter 102 would be always equal to η0 irrespective of the input voltage of the main signal A, as shown in FIG. 2 by line c22. Unfortunately, it is extremely difficult in practice to have a power efficiency of the Chireix transmitter 102 exceeding the efficient of the Chireix transmitter 101 by enhancing the power conversion efficiency of the high frequency-DC converter circuit 172. Therefore, so far as the present inventor knows, the Chireix transmitter 102 has not been in actual practice, though its value has been studied in some academic fields.
Next, a Doherty transmitter 103 will now be discussed, which is one of variable-load impedance type composite transmitters equally well known as the Chireix transmitter. Referring to FIG. 4(a), there is shown a Doherty transmitter 103, which comprises:
(1) an orthogonal modulator 90 which, upon receipt of an in-phase baseband signal I(t) and an orthogonal baseband signal Q(t), outputs a main signal A obtained by orthogonal modulation of the two input signals;
(2) a power amplifier 152 for power amplifying the main signal A fed thereto;
(3) a power amplifier 153 for power amplifying the main signal A fed thereto after the main signal A is delayed by means of a ¼-wavelength line 163; and
(4) a Doherty power combiner 142 for combining the outputs of the power amplifiers 152 and 153 via an impedance inverter 162 to thereby provide a transmission output signal S0.
In the Doherty transmitter 103, the power amplifier 152 is called a carrier amplifier (CA), which is in practice either B- or AB-class amplifier. The power amplifier 153, which is also called peaking amplifier (PA), is a C-class CA. Each of the CA and the PA receives a bifurcated and power-amplified main signal A. The Doherty transmitter 103 can be represented by an ideal current source model as shown in FIG. 4(b), assuming that the ¼-wavelength line functions as an ideal impedance inverter 162 and that the CA and the PA are both ideal current sources. FIG. 5(a) shows a normalized output voltage (characteristic curve c50) of the CA as a function of the normalized input voltage of the main signal, along with a normalized output voltage of the PA (characteristic curve c51). FIG. 5(b) shows normalized output current characteristics of the CA and PA (curves c52 and c53) as functions of the normalized input voltage of the main-signal.
Operations of the Doherty transmitter 103 may be discussed in two separate domains, one in a small power domain (where the envelope level is not more than ½ of the peak vale C of the signal) and in a large power domain (where the envelope level of the main signal exceeds ½ of the peak level.) Referring to curve c50 shown in FIG. 5(a) and curve c53 shown in FIG. 5(b), it is seen that the CA operates but the PA is cut off in the small power domain, so that the PA can be regarded as an open circuit. Consequently, if the loaded impedance of the CA (assumed to be a B-class amplifier) is 50Ω, it operates as an ordinary B-class amplifier, supplying power to a 100Ω load. Its instantaneous power efficiency increases with the output voltage, and reaches its 78.5% efficiency at the normalized main signal voltage of 0.5.
As the voltage of the main signal exceeds one half the peak envelope level C, the PA begins to operate, causing a further current to be supplied from the PA to the Doherty transmitter 103, thereby reducing its apparent load impedance. As the CA remains saturated at a saturation point, maintaining a constant voltage, the CA can be regarded as a constant-voltage power source that operates at its maximum power efficiency. When outputting a peak envelope power (PEP), the CA and the PA can see a load of 50Ω, each providing its power as much as ½ of the maximum output power, and then the theoretical PEP efficiency of the CA is 78.5% if the CA is a B-class amplifier (Non-patent Document 6).
It has been disclosed in a literature (Non-patent Document 6) that when the Doherty transmitter 103 shown in FIG. 4(b) is represented by an ideal current source, its power efficiency is represented by curve c60 shown in FIG. 6 (to be compared with curve c61 representing the power efficiency of a B-class amplifier). In actuality, however, it is difficult to make the Doherty transmitter 103 operable over a wide frequency domain due to the fact that in a domain away from its central frequency fc its power efficiency drops and causes its output signal distorted, although the ¼-wavelength line 162 acts as an ideal impedance inverter at the central frequency fc (Patent Document 1).
In recent years, it has been proposed in Non-patent Document 6 (FIG. 7(a)) to amplitude modulate the main signal A in such a way that the input voltage vs output current characteristic of the power amplifier 152 becomes identical to the input voltage vs output power characteristic of the CA used in the Doherty transmitter 103 and the input voltage vs output current characteristic of the power amplifier 153 becomes identical to the input voltage vs output power characteristic of the PA utilized in the Doherty transmitter 103 when the power amplifiers 152 and 153 are of B-class or AB-class amplifiers.
A merit of the Doherty transmitter 104 lies in the fact that the input power fed to the power amplifiers 152 and 153 can be reduced as compared with the Doherty transmitter 103, which helps increase the power add efficiency of the Doherty transmitter 104, and that the same B- or AB-class amplifiers can be used as the two power amplifiers 152 and 153, which permits reduction of its manufacturing cost and extension of operable frequency band, as disclosed in the Patent Document 1.
The EIS to be added to the main signal A of the Doherty transmitter 104 will be referred to as EIS A1, as in the Chireix transmitter 101. By writing the main signal A in the form of Eq. 1 and Eq. 2 as in the Chireix transmitter 101, the EIS A1 is given by
                                                                        A                ⁢                                                                  ⁢                1                ⁢                                  (                  t                  )                                            =                            ⁢                                                a                  ⁡                                      (                    t                    )                                                  ⁢                cos                ⁢                                  {                                                                                    ω                        0                                            ⁢                      t                                        +                                          φ                      ⁡                                              (                        t                        )                                                                              }                                                                                                      ⁢                              (                                  0                  ≤                                      a                    ⁡                                          (                      t                      )                                                        ≤                                      C                    /                    2                                                  )                                                                                        =                            ⁢                                                {                                      C                    -                                          a                      ⁡                                              (                        t                        )                                                                              }                                ⁢                cos                ⁢                                  {                                                                                    ω                        0                                            ⁢                      t                                        +                                          φ                      ⁡                                              (                        t                        )                                                                              }                                                                                                      ⁢                              (                                                      C                    /                    2                                    ≤                                      a                    ⁡                                          (                      t                      )                                                        ≤                  C                                )                                                                        (                  Eq          .                                          ⁢          10                )            The first composite signal S1(t) fed to the power amplifier 152 is a vectorial sum of the main signal A and the EIS A1, while the second composite signal S2(t) fed to the power amplifier 153 is a vectorial difference between the main signal A and the EIS A1, which are give by the following equations, respectively
                                                                        S                ⁢                                                                  ⁢                1                ⁢                                  (                  t                  )                                            =                            ⁢                                                A                  ⁡                                      (                    t                    )                                                  +                                  A                  ⁢                                                                          ⁢                  1                  ⁢                                      (                    t                    )                                                                                                                                                                                    =                            ⁢                              2                ⁢                                  a                  ⁡                                      (                    t                    )                                                  ⁢                cos                ⁢                                  {                                                                                    ω                        0                                            ⁢                      t                                        +                                          φ                      ⁡                                              (                        t                        )                                                                              }                                                                                                      ⁢                              (                                  0                  ≤                                      a                    ⁡                                          (                      t                      )                                                        ≤                                      C                    /                    2                                                  )                                                                                        =                            ⁢                              C                ⁢                                                                  ⁢                cos                ⁢                                  {                                                                                    ω                        0                                            ⁢                      t                                        +                                          φ                      ⁡                                              (                        t                        )                                                                              }                                                                                                      ⁢                              (                                                      C                    /                    2                                    ≤                                      a                    ⁡                                          (                      t                      )                                                        ≤                  C                                )                                                                        (                  Eq          .                                          ⁢          11                )                                                                                    S                ⁢                                                                  ⁢                2                ⁢                                  (                  t                  )                                            =                            ⁢                                                A                  ⁡                                      (                    t                    )                                                  -                                  A                  ⁢                                                                          ⁢                  1                  ⁢                                      (                    t                    )                                                                                                                                                                                    =                            ⁢              0                                                                        ⁢                              (                                  0                  ≤                                      a                    ⁡                                          (                      t                      )                                                        ≤                                      C                    /                    2                                                  )                                                                                        =                            ⁢                                                {                                                            2                      ⁢                                              a                        ⁡                                                  (                          t                          )                                                                                      -                    C                                    }                                ⁢                cos                ⁢                                  {                                                                                    ω                        0                                            ⁢                      t                                        +                                          φ                      ⁡                                              (                        t                        )                                                                              }                                                                                                      ⁢                              (                                                      C                    /                    2                                    ≤                                      a                    ⁡                                          (                      t                      )                                                        ≤                  C                                )                                                                        (                  Eq          .                                          ⁢          12                )            
Patent Document 1 proposes to replace, by an alternative Doherty transmitter 105 (not shown), a circuit of the Dohery transmitter 104 that generates a first and a second composite signals S1 and S2, respectively, from a main signal. The Doherty transmitter 105 generates the second composite signals S2 by passing the main signal A through a nonlinear circuit (nonlinear emulator 181), and then generates the first composite signal S1 by passing S2 through a cross combining filter 182 and subtracting the resultant signal from the main signal A.
In a conventional Doherty transmitter 103, the input vs output voltage characteristic of the CA for inputted main signal is not constant in a large-power domain, while the input vs output voltage characteristic of the PA lacks linearity in a small-power domain, as shown in FIG. 8(a) by curves c80 and c81, respectively. On the other hand, in an attempt to broaden such linearity domain, Patent Document 1 discloses a method of simulating ideal input-output voltage characteristics of the CA and PA for a main signal as shown in FIG. 8(b) (curves C82 and C83) by optimization of a cross coupling filter 182.
In any of conventional composite transmitters discussed above, a usable frequency bandwidth of the EIS is limited by a limitative operational speed of a digital signal processing circuit used. Furthermore, in actuality the first nor the second power amplifiers can never be a perfect linear amplifier, outputting nonlinear distorted components even outside the limited frequency band of the EIS.
This phenomenon is called “spectral regrowth”, which will be now described with reference to Fig. (a)-(b) for a case where the first and the second power amplifiers are of AB-class (having a gate bias voltage 0.2 times that of an A-class amplifier), and a main signal is an standard RF modulated signal (or 4-wave QPSK signal, as defined and used in the first embodiment below).
FIG. 9(a) shows power spectral densities (PSDs) of the EIS (curve c91) and of the output signal (curve c92) of the Chireix transmitter 101, showing spectral regrowths outside the normalized frequency band (−1.5 through +1.5).
FIG. 9(b) shows power spectral densities of the EIS (curve c94) and of the output signal (curve c94) of the Doherty transmitter 104, showing spectral regrowths outside the normalized frequency band (−1.5 through +1.5). In view of the fact that a spectral regrowth density decreases as the PSD decreases at a frequency band edge of the EIS, it is an important issue for a composite transmitter to narrow its EIS bandwidth for an improvement of the ACLR characteristic of the composite transmitter.